Sensor arrangement and method for operating a sensor arrangement

ABSTRACT

A sensor arrangement and method of operating the sensor arrangement are described. The sensor arrangement has sensor devices formed on or in a substrate and a calibration device for calibrating a respective sensor device. Each sensor device has an electrical signal converter which is a field-effect transistor (FET) having a gate terminal coupled to a sensor element that is coupled to the signal converter, a device that keeps constant the electrical voltage between source and drain terminals of the FET, and a device for detecting the value of the electric current flowing through the signal converter. The sensor element characteristically influences the electrical conductivity of the signal converter on account of a sensor event on the sensor element. The calibration device brings the gate region of the FET to an electrical calibration potential such that the electric current is independent of parameter fluctuations of the FET.

The invention relates to a sensor arrangement and a method for operatinga sensor arrangement.

The interface between biology and semiconductor technology is ofinterest from scientific and economic standpoints. Mention may be madein this connection of, by way of example, the coupling betweenbiological cell assemblages, such as neurons for example, and siliconmicroelectronics. A biological system may be examined in spatially ortemporally resolved fashion on the surface of asemiconductor-technological sensor by means of sensor elements arrangedin matrix form. In particular, metabolism parameters of the cells can berecorded for example by detecting local pH values with the aid ofion-sensitive field-effect transistors (ISFETs) as sensor elements. AnISFET has an ion-sensitive layer in operative contact with electricallycharged particles to be detected, the electrically charged particlescharacteristically influencing the conductivity of the ISFET, which canbe detected as a sensor variable.

Examining the reaction of a biological system to an electricalstimulation is of great interest. Neurons (nerve cells) can generate asmall electric current via ion channels in the cell membranes inspecific regions of their surface, said current being detected by asensor situated underneath. The requisite stringent requirements made ofspatial and temporal resolution of the sensor are achieved by means ofsilicon microelectronics.

Matrix-type arrangements of sensor elements are also used in otherimportant fields such as medical analysis or DNA sensor technology.

For these and other possible applications of semiconductor sensors inintegrated circuits, it is advantageous to integrate a large number ofsensors in a common sensor array. Even in the case of small sensorarrays (and all the more so in the case of large sensor arrays), notevery sensor can be connected to a dedicated read-out circuit. Thenumber of requisite lines and also the redundancy of the read-outcircuits arranged outside the array are factors opposing such asolution.

FIG. 1 schematically shows a sensor arrangement 100 in accordance withthe prior art having four sensor devices 101 to 104, a multiplexerrepresented as switch 105, and a read-out circuit 106.

The outputs of the individual sensor devices are read out using themultiplexer 105, that is to say serially. The respective sensor device101 to 104 coupled to the read-out circuit 106 via the multiplexer 105outputs an output voltage correlated with the event to be detected (forexample the illumination intensity in the case of a photosensor or anelectrical signal of a nerve cell arranged on the respective sensordevice). In order to represent these facts schematically, the sensordevices 101 to 104 are represented schematically as voltage sources V₁,V₂, V₃, V₄.

In accordance with the scenario illustrated in FIG. 1, the first sensordevice 101 is coupled to the read-out circuit 106 via the multiplexer105. By means of the read-out circuit 106, the output voltage of arespective sensor device 101 to 104 is amplified for further processing.

If it is necessary to record the measured values of a multiplicity ofsensor devices with a high sampling rate, then an individual sensordevice has to be read in a very short time. This results in a stringentrequirement made of the temporal resolution of the system.

The sensor arrangement 200 in accordance with the prior art as shown inFIG. 2 exhibits, in addition to the components described in FIG. 1, aninternal resistance R_(i) 201 of each of the sensor devices and also acommon capacitance C 202 of the sensor devices 101 to 104. The timeconstant of the sensor arrangement 200 is determined by means of the RCelement comprising internal resistance R_(i) of the sensor devices 101to 104 and also the capacitance C 202 that is to be subjected to chargereversal. These two parameters are subject to specific limitations for aspecific production technology. Consequently, the achievable timeconstant τ=RC is also limited.

The time constant τ clearly specifies the time after which the outputsignal has risen to (1−1/e)=63% of the final value. If measurementerrors caused by the transient response of the sensor arrangement 200are intended to lie below a specific, just still tolerable limit, it isnecessary, under certain circumstances, to wait during multiples ofthese time constants (e.g. 2τ or 3τ). Consequently, the time constantτ=RC is a characteristic measure of the minimum time that can beattained between two successive measurements. Therefore, the value of τresults in a limitation of the maximum number of sensor devices that canbe read within a predetermined time segment at a predetermined samplingrate.

FIG. 3 is a diagram 300 schematically showing the time dependence of thesignal profile at a sensor device 201 to 204 after the production of thecoupling (t=0) of the respective sensor device to the read-out circuit106 via the multiplexer 105.

The time that has elapsed since the changeover of the multiplexer 105(t=0) is plotted along the abscissa 301 of the diagram 300. At theinstant t₀, the signal profile has for the first time fallen below thepredetermined tolerable value V_(tol). The temporal signal profile atthe voltage sources V₁ to V₄ represented schematically in FIG. 1, FIG.2, is represented along the ordinate 302 of the diagram 300. The signalprofile curve 303 reflects the transient response at the capacitance 202and the resistances R_(i), and to a good approximation is a fallingexponential function. Clearly, at the instant t₀, the dynamic error hasfor the first time fallen below the tolerable error, so that themeasurement time that is necessary at the least is t₀.

In the case of sensor arrangements disclosed in the prior art, thesensor devices of a sensor array are read in the voltage domain, that isto say that the measurement variable is converted into an electricalvoltage. In order to keep down the RC constant, and therefore to enablesufficiently many sensor devices to be read sufficiently rapidly,attempts are made to reduce the internal resistance R_(i) and, as aconsequence thereof, the time constant τ.

One possibility for reducing the internal resistance R_(i) of the sensordevices independently of the circuit architecture is to increase thedriver capability of the output transistors of a circuit. Thedisadvantage of this measure is an increased area requirement and agreater capacitive loading on the preceding amplifier stage or theconnected sensor, which leads to an attenuation of the signal. Onaccount of the frequently severe area limitation and the need forefficient area utilization, this solution is not suitable for manyfields of application.

Usually, for integrated circuits operated in the voltage domain, aseries of basic circuits are used for the output drivers. What isdisadvantageous about an additional output driver circuit integratedinto the sensor devices is the increase in the space requirement of asensor device and, consequently, the reduction of the achievable spatialresolution. A boundary condition for the consideration below is that anoutput transistor coupled to a sensor electrode is simultaneously usedfor signal amplification. This is not a mandatory prerequisite, however.

FIG. 4A illustrates an output driver circuit 400, in the case of which aMOS transistor 401 is operated in a common-source connection.

The sensor device is represented as voltage source V_(G) 402.Furthermore, the output driver circuit 400 is shown with aconstant-current source I₀ 403, a nonreactive resistance R 404representing the internal resistance of the sensor device, and also acapacitance C 405 representing the capacitance of a sensor arrangement.The output voltage V_(out) is present across the capacitance 405. Thevoltage source 402 is coupled to the gate terminal of a MOS transistor401. One source/drain terminal of the MOS transistor 401 is at groundpotential 407, whereas the other source/drain terminal of the MOStransistor 401 is coupled both to the constant-current source 403 and tothe nonreactive resistance 404 and to the capacitance 405. The operatingpoint of the MOS transistor 401 is determined by means of theconstant-current source 403, the nonreactive resistance 404 and thevoltage source V_(G) 402.

FIG. 4B shows a small-signal equivalent circuit diagram 410 of theoutput driver circuit 400 shown in FIG. 4A. The equivalent circuitdiagram 410 shows a controlled current source 411 (g_(m)ΔV_(G)), aneffective internal resistance 412 (g_(ds)+R⁻¹)⁻¹ and also thecapacitance 405 that is to be subjected to charge reversal. g_(ds)designates the output conductance of the MOS transistor 401, and g_(m)is the transconductance of the MOS transistor 401.

In the case of the common-source connection of the MOS transistor 401 asshown in FIG. 4A, FIG. 4B, one source/drain terminal is at groundpotential 407. The operating point of the output driver circuit 400 ispredetermined by means of the DC component of the voltage source V_(G)402 at the gate terminal of the MOS transistor 401 and by means of theconstant-current source I₀ with the internal resistance R coupled to theother source/drain terminal of the MOS transistor 401. A solutionwithout a constant-current source I₀ 403, just with a nonreactiveresistance R 404, is also possible as an alternative. In the case of theoutput driver circuit 400 shown in FIG. 4B, a supply voltage 406 isapplied to a respective terminal of the constant-current source 403 andof the nonreactive resistance 404, whereas a terminal of the voltagesource 402, one source/drain terminal of the MOS transistor 401 and aterminal of the capacitance 405 are at ground potential 407.

If the gate voltage V_(G) is modulated (for example on account of asensor event), then the drain current ID of the MOS transistor 401changes by g_(m)ΔV_(G). This is symbolized by means of the controlledcurrent source 411 from FIG. 4B, showing the small-signal equivalentcircuit diagram of the output driver circuit 400. The change in theoutput voltage ΔV_(out) results from the changed voltage drop across theMOS transistor 401 and across the effective resistance 412. For lowfrequencies ω, the capacitance C can be disregarded to a goodapproximation, and this results in an open-loop gain A(ω=0) which can bedescribed by means of the following expression:A(ω=0)=ΔV _(out) /ΔV _(G) =g _(m)/(g _(ds) +R ⁻¹)   (1)

If transistor parameters that are typical of CMOS technology areinserted into equation (1), this results in a possible voltage gain by afactor of approximately ten to approximately fifty. As a result, evensignals having a small amplitude are amplified after the first amplifierstage such that they are no longer appreciably disturbed by noiseeffects. The time constant τ_(c) of the amplifier results as:τ_(c) =C/(g _(ds) +R ⁻¹)   (2)

The capacitance is predetermined by the technology used and, in thisrespect, can only be influenced to a very limited extent, for example bymeans of optimizing the layout. The parameters R and g_(ds) cannot bevaried arbitrarily for a specific gain and on account of arealimitations, so that a limit value for the maximum bandwidth that can betransmitted results from these boundary conditions.

A description is given below, referring to FIG. 5A, of a further outputdriver circuit 500 in accordance with the prior art. An equivalentcircuit diagram 510 of the output driver circuit 500 shown in FIG. 5A isdescribed referring to FIG. 5B. Those components of the output drivercircuit 500 which are also contained in the output driver circuit 400are provided with the same reference numerals.

In the case of the output driver circuit 500 shown in FIG. 5A, the MOStransistor 401 is configured in a source follower connection. In adeparture from the output driver circuit 400 shown in FIG. 4A, thatsource/drain terminal of the MOS transistor 401 which is not coupled tothe constant-current source 403, the voltage source 402 and thecapacitance 405 is at the potential of the supply voltage 406.Furthermore, a terminal of the constant-current source 403 and aterminal of the capacitance 405 are at the electrical ground potential407 in the case of the output driver circuit 500. The constant-currentsource I₀ 403 and the voltage source V_(G) 402 determine the operatingpoint of the MOS transistor 401.

The electric current at the source/drain terminal of the MOS transistor401 that is coupled to the constant-current source I₀ 403 is determinedby means of the constant-current source I₀ 403. The gate voltage of theMOS transistor 401 is set by means of the voltage source V_(G) 402.Furthermore, a capacitance C 405 that is to be subjected to chargereversal is shown.

The small-signal equivalent circuit diagram 510 of FIG. 5B shows acontrolled current source g_(m)(ΔV_(G)-ΔV_(out)) 511 and also aninternal resistance g_(ds). In accordance with the small-signalequivalent circuit diagram 510 shown in FIG. 5B, the field-effect-basedmodulation of the electric current at the lower source/drain terminal ofthe MOS transistor 401 in accordance with FIG. 5A is represented ascontrolled current source g_(m)(ΔV_(G)-V_(out)) 511.

The dependence of said electric current on the electrical voltage at thelower source/drain terminal of the MOS transistor 401 in accordance withFIG. 5A is determined by the conductance g_(ds). The time constant τ_(c)of the amplifier is determined by the RC element and is described by thefollowing expression:τ_(c) =C/(g _(ds) +g _(m))   (3)

The time constant τ_(c) from equation (3) corresponds to the scenario ofthe common-source connection of the MOS transistor 401 from FIG. 4A ifthe value of the nonreactive resistance R is equal to the inversetransconductance g_(m) ⁻¹ of the MOS transistor 401 (cf. equation (2)).In accordance with the two circuit architectures, the gain is thenapproximately one, namely:A(ω=0)=ΔV _(out) /ΔV _(G) =g _(m)/(g _(m) +g _(ds))≦1   (4)

This low gain is disadvantageous, on account of the small signalamplitudes that have to be conducted out from the sensor device, sincenoise effects can corrupt the measured signal.

As an example of a realization of the principle described on the basisof a CMOS camera, reference shall be made to [1].

To summarize, the functionality of the circuit architectures disclosedin the prior art for reading out sensor signals of a sensor array isinadequate since a large time constant for reading the individual sensordevices results from the capacitance that is to be subjected to chargereversal. This leads to a poor temporal resolution. Furthermore, thegain of the often small sensor signal is often insufficient in the caseof circuit architectures disclosed in the prior art.

The invention is based on the problem of providing a sensor arrangementhaving high spatial resolution which, in conjunction with a sufficientlyhigh signal gain, enables a fast read-out of sensor signals, i.e. a highbandwidth.

The problem is solved by means of a sensor arrangement and by means of amethod for operating a sensor arrangement having the features inaccordance with the independent patent claims.

The sensor arrangement according to the invention contains a pluralityof sensor devices formed on and/or in a substrate. Each sensor devicehas an electrical signal converter and a sensor element coupled to thesignal converter, which sensor element can be used to characteristicallyinfluence the electrical conductivity of the signal converter on accountof a sensor event on the sensor element. Furthermore, the sensor deviceaccording to the invention has a device for keeping constant anelectrical voltage present at the signal converter. Moreover, eachsensor device has a device for detecting the value of the electriccurrent flowing through the signal converter as sensor signal.

Furthermore, the invention provides a method for operating a sensorarrangement having the features mentioned above, in which case, inaccordance with the method, the electrical conductivity of the signalconverter is characteristically influenced on account of a sensor eventon a respective sensor element. Furthermore, the electrical voltage atthe signal converter is kept constant. The electric current flowingthrough the signal converter is detected as sensor signal.

A fundamental idea of the invention is based on detecting, instead ofthe electrical voltage, an electric current at a signal convertercoupled to the sensor element. By virtue of an electric current beingdetected according to the invention, rather than an electrical voltagein accordance with the prior art, a charge reversal of capacitances isavoided. A larger bandwidth, that is to say a faster read-out of thesensor elements of a sensor arrangement, is made possible as a result.The time constant of the system is no longer defined by theinterconnection within the sensor device, but rather only by theexternal circuit. Clearly, the electrical voltage as sensor signal isstabilized externally according to the invention. With the circuitarchitecture according to the invention for a sensor arrangement, aparticularly high number of sensor elements and a particularly highsampling rate are made possible for a predetermined technology, therebyachieving a small time constant for reading the sensor elements. Onaccount of the interconnection of the sensor device according to theinvention, an electric sensor current is detected instead of anelectrical sensor voltage, which leads to a high gain and a small timedelay.

If, instead of the electrical voltage V_(out), the electric currentI_(out) is used as output variable of the output stage of a sensordevice or of the signal converter, then the voltage across thecapacitance becomes independent of the output variable and can be keptconstant at a value of the output voltage, by way of example. Since theinductive properties of integrated leads are generally negligiblecompared with the capacitive properties, this results in a considerablyreduced time constant. In practice, an unavoidable internal resistanceof the measuring circuit leads to a small voltage drop. However, sincethe measuring circuit is situated outside the sensor devices, itsinternal resistance can be kept down. As a result, the time constantτ=RC is orders of magnitude smaller than in accordance with the priorart. Sensor arrangements that can be read faster or an increased numberof sensor devices are made possible as a result. Clearly, an outputdriver circuit of a sensor device is embodied in a “current modetechnology”.

A further important aspect of the invention is to be seen in the factthat a plurality of sensor devices are formed on and/or in thesubstrate. In other words, the sensor arrangement according to theinvention is realized as an integrated circuit, for example in and/or ona silicon substrate (e.g. wafer, chip, etc.). Miniaturization isachieved as a result and an array having a high number of sensor devicesis thus produced. Furthermore, the sensor arrangement can be fabricatedwith tenable outlay using the modern and mature silicon microtechnology.

Preferred developments of the invention emerge from the dependentclaims.

In the case of the sensor arrangement, the electronic signal converteris preferably a transistor (e.g. a bipolar transistor).

In at least some of the sensor devices according to the invention, theelectronic signal converter may be a field-effect transistor whose gateterminal is coupled to the sensor element, the device for keepingconstant an electrical voltage being set up in such a way that it keepsconstant the electrical voltage between the source/drain terminals ofthe field-effect transistor. A field-effect transistor as signalconverter has the functionality that a modulation of the gate voltage isconverted into an altered electrical conductivity of the channel regionof the field-effect transistor, so that the value of the electriccurrent flowing through a source/drain terminal is characteristicallyinfluenced on account of the altered nonreactive resistance of thechannel region of the field-effect transistor.

As an alternative to a field-effect transistor, the electronic signalconverter may be embodied as an arbitrarily configured controllablenonreactive resistor, for example as a potentiometer, the resistance ofwhich is controlled for example by means of an electrical signal onaccount of a sensor event.

Furthermore, the sensor device according to the invention may have anevaluation unit, the evaluation unit being provided with the value ofthe electric current as sensor signal.

The evaluation unit is preferably set up in such a way that it forms,from the value of the electric current, an electrical voltagecharacteristic of this value or maps the value of the electric currentonto a digitally coded value that characterizes the latter.

In other words, the detected electric current can be converted into anelectrical voltage, which may be advantageous for the further processingof the signal. Furthermore, an analog current signal can be convertedinto a digital signal—and hence a signal that is more robust in respectof errors.

In particular, the evaluation unit may have an operational amplifier, inparticular connected up as a voltage follower, having a first input, towhich the sensor signal can be applied. Furthermore, the operationalamplifier has a second input, to which an electrical reference potentialcan be applied. The characteristic electrical voltage is provided at anoutput of the operational amplifier, the first input and the outputbeing coupled to one another by means of a nonreactive resistor.

The sensor arrangement may be configured as a biosensor arrangement. Thesensor element of each sensor device may for example detect anelectrical signal of a nerve cell grown on the sensor element. As analternative, the sensor element may detect electrically chargedparticles on the sensor element-using an ISFET.

The sensor arrangement may have a calibration device for calibrating arespective sensor device, which is set up in such a way that it can beused to bring the gate region of the field-effect transistor to anelectrical calibration potential such that the electric current isindependent of parameter fluctuations of the field-effect transistor. Byway of example, field-effect transistors of different sensor devices mayhave different parameters (e.g. threshold voltage) due to a fabricationmethod. The calibration makes it possible to ensure that parameterfluctuations do not lead to corruption in the detection of a sensorevent.

In particular, the sensor device can be calibrated by means ofimplementing a calibration device based on the autozeroing technique,thereby providing a robust sensor arrangement. A signal influencing onaccount of parameter fluctuations of the components of a respectivesensor device, for example the signal converters realized asfield-effect transistors, is thereby avoided.

The calibration device may be set up in such a way that an electriccalibration current can be applied to the gate terminal and to asource/drain terminal of the field-effect transistor for calibrationpurposes.

As an alternative, the evaluation unit may have a correlated doublesampling device, which may be set up in such a way that it forms, in thecase of a sensor event, a value of the electric current that isindependent of parameter fluctuations of a respective field-effecttransistor.

In accordance with the correlated double sampling principle, clearly ina first step a sensor event is detected in a sensor device and thesensor signal is stored. This sensor signal is dependent on thealteration of the value of the physical parameters of the sensor device(for example geometrical parameters of a field-effect transistor) andmay furthermore be dependent on physical parameters of furthercomponents, for example an amplifier for amplifying the sensor signal.In a second step, an auxiliary signal that depends only on the value ofthe physical parameter of the sensor device is detected in the absenceof a sensor event. If the auxiliary signal is subtracted from the sensorsignal, then a sensor signal that is essentially independent of thevalue of the physical parameter is obtained.

In particular, the correlated double sampling device of the inventionmay be set up in such a way that, by means of this device, in acalibration phase, the gate region of the field-effect transistor isbrought to an electrical calibration potential and the associated valueof the electric current is detected as calibration signal and stored. Ina detection phase, the value of the electric current on account of asensor event is detected as sensor signal. In an evaluation phase,sensor signal and calibration signal can be evaluated jointly.

Preferably, the sensor devices of the sensor arrangement are arrangedessentially in matrix form on and/or in the substrate and are connectedup by means of row and column lines in such a way that the sensordevices can be driven individually, row by row or column by column.

In the case of the sensor arrangement according to the invention, atleast one evaluation unit, at least one calibration device and/or atleast one correlated double sampling device may be provided jointly forat least a portion of the sensor devices of a row line or a column line.

The method according to the invention for operating the sensorarrangement according to the invention is described in more detailbelow. Refinements of the sensor arrangement are also applicable to themethod for operating the sensor arrangement, and vice versa.

In accordance with the method according to the invention, a field-effecttransistor whose gate terminal is coupled to the sensor element may beused as the electronic signal converter of a respective sensorarrangement, the electrical voltage between the source/drain terminalsof the field-effect transistor being kept constant by means of thedevice for keeping constant an electrical voltage.

Furthermore, a respective sensor device may be calibrated by the gateregion of the field-effect transistor being brought to an electricalcalibration potential such that the value of the electric current in thecase of a sensor event becomes independent of the properties of thefield-effect transistor (e.g. a production-dictated deviation of thethickness of the gate insulating layer from a desired or average value).

A value of the electric current that is independent of the properties ofthe field-effect transistor may be formed using the correlated doublesampling method in the case of a sensor event.

Exemplary embodiments of the invention are illustrated in the figuresand are explained in more detail below.

In the figures:

FIG. 1 shows an illustration of one sensor arrangement in accordancewith the prior art,

FIG. 2 shows an illustration of another sensor arrangement in accordancewith the prior art,

FIG. 3 shows a diagram illustrating a signal profile as a function of aread-out time,

FIG. 4A, FIG. 4B show one output driver circuit and an associatedsmall-signal equivalent circuit diagram in accordance with the priorart,

FIG. 5A, FIG. 5B show another output driver circuit and an associatedsmall-signal equivalent circuit diagram in accordance with the priorart,

FIG. 6 shows a sensor device in accordance with a preferred exemplaryembodiment of the invention,

FIG. 7 shows an evaluation unit according to the invention,

FIG. 8 shows a sensor arrangement in accordance with a first exemplaryembodiment of the invention,

FIG. 9 shows a sensor arrangement in accordance with a second exemplaryembodiment of the invention,

FIG. 10 shows a sensor arrangement in accordance with a third exemplaryembodiment of the invention.

A description is given below, referring to FIG. 6, of a sensor device600 in accordance with a preferred exemplary embodiment of theinvention.

The sensor device 600 shown in FIG. 6 has a field-effect transistor 601as electrical signal converter. Furthermore, the sensor device 600 has abiosensor element coupled to the field-effect transistor 601, whichbiosensor element is represented schematically as voltage source 602 inFIG. 6. If a sensor event takes place at the biosensor element, then theelectrical conductivity of the channel region of the field-effecttransistor 601 is characteristically influenced on account of saidsensor event.

A first source/drain terminal 601 a of the field-effect transistor 601is coupled to one terminal of an ammeter 603 for detecting a sensorcurrent I_(out), a supply voltage 604 being applied to the otherterminal of said ammeter. A second source/drain terminal 601 b of thefield-effect transistor 601 is at the ground potential 605. On accountof this interconnection, a constant electrical potential difference isapplied between the source/drain terminals 601 a, 601 b. Furthermore,the constant voltage V_(out), which results from the difference betweenthe supply voltage 604 and the ground potential 605, is also presentacross a capacitor 606 representing the effective capacitance of thesensor device 600. The ammeter 603 is a device for detecting the valueof the electric current flowing through the first source/drain terminal601 a of the field-effect transistor 601 as sensor signal.

The gate terminal 601 c of the field-effect transistor 601 is coupled tothe voltage source 602. The electrical voltage between the source/drainterminals 601 a, 601 b of the field-effect transistor 601 is constant.The first source/drain terminal 601 a of the field-effect transistor 601is coupled to one terminal of the capacitor 606 and is furthermorecoupled to the ammeter 603.

If a sensor event takes place on the biosensor element, then theelectrical voltage V_(G) of the voltage source 602 is thereby modulated.The latter is provided at the gate terminal 601 c of the field-effecttransistor 601, so that the electrical conductivity of the channelregion of the field-effect transistor 601 is thereby characteristicallyinfluenced. As a result, the value of the electric current flow throughthe source/drain terminal 601 a of the field-effect transistor 601,which current flow is detected by the ammeter 603, is a characteristicmeasure of the sensor event.

The field-effect transistor 601 is operated in the “current mode”. Thevoltage source 602 V_(G) determines the potential present at the gateregion 601 c, whereas the voltage at the first source/drain terminal 601a is fixed at the supply voltage 604. The output current I_(out) isdetected by means of the ammeter 603. It should be noted that thecapacitance C 606 does not have to be subjected to charge reversal sincethe voltage is kept constant.

A description is given below, referring to FIG. 7, of an evaluation unit700 in accordance with a preferred exemplary embodiment of theinvention.

The evaluation unit has the functionality of keeping constant theelectrical voltage on a sensor line and of simultaneously converting thedetected sensor current (preferably linearly) into a sensor voltage.This enables further processing in the voltage domain.

For some applications it may be advantageous for a detected electricsensor current to be converted into a voltage signal. The evaluationunit 700 shown in FIG. 7 shows an exemplary embodiment that enables acurrent signal ΔI_(meas) to be converted into a voltage signal ΔV_(out).The circuit shown in FIG. 7 is used both to keep constant the electricalvoltage on the line and to convert the modulated current I_(meas) into amodulated output voltage V_(out)=I_(meas)Z. In this case, Z denotes thevalue of an impedance 701.

As shown in FIG. 7, the modulated value of the electric currentΔI_(meas) is provided as sensor signal at an input 702 of the evaluationunit 700. The evaluation unit 700 has an operational amplifier 703. Theoperational amplifier 703 has a noninverting input 703 a, at which thesensor signal ΔI_(meas) is provided. Furthermore, the operationalamplifier 703 has a noninverting input 703 b, to which a constantelectrical reference potential V_(kal) is applied. The characteristicelectrical voltage ΔV_(out) is provided at an output 703 c of theoperational amplifier 703. The output 703 c of the operational amplifier703 is fed back to the inverting input 703 a of the operationalamplifier 703 via the impedance Z 701. As is furthermore shown in FIG.7, the electrical potential V_(kal) at the noninverting input 703 b ofthe operational amplifier 703 is provided by means of theconstant-voltage source 704. The constant-voltage source 704 isconnected between the electrical ground potential 605 and thenoninverting input 703 b of the operational amplifier 703.

A description is given below, referring to FIG. 8, of a sensorarrangement 800 in accordance with a first preferred exemplaryembodiment of the invention.

The sensor arrangement 800 has a multiplicity of sensor devices 801which are arranged in matrix form and are connected up by means of rowand column lines in such a way that the sensor devices 801 can be drivenindividually or column by column.

The sensor devices 801 of the sensor arrangement 800 are formed on andin a silicon substrate (not shown). In other words, the sensorarrangement 800 is realized as an integrated circuit.

Although the sensor arrangement 800 has a multiplicity of sensor devices801, only the n-th and (n+1)-th column of sensor devices 801 and alsothe m-th and (m+1)-th row of sensor devices 801 are shown in FIG. 8 forthe purpose of a simplified illustration.

The construction of the sensor device 801 arranged in the n-th columnand the m-th row from FIG. 8 is described in more detail by way ofexample below.

A sensor element 801 of the sensor arrangement 800 is shownschematically as voltage source 802 in FIG. 8, said sensor element beingarranged between the ground potential 605 and one terminal of acapacitor 803. The capacitor 803 symbolizes a dielectric which isapplied on the sensor element and spatially decouples electricallycharged particles to be detected that are situated thereon from the gateregion 804 c of a detection transistor 804. The other terminal of thecapacitor 803 is furthermore coupled to the first source/drain terminal805 a of a calibration transistor 805. The second source/drain terminal805 b of the calibration transistor 805 is coupled to the firstsource/drain terminal 804 a of the detection transistor 804.Furthermore, the first source/drain terminal 804 a of the detectiontransistor 804 is coupled to the first source/drain terminal 806 a of aselection transistor 806. The second source/drain terminal 806 b of theselection transistor 806 is coupled to a changeover element 807, whichis provided jointly for each row of sensor devices 801. The gateterminal 806 c of the selection transistor 806 is coupled to a firstcolumn line 808, which is provided jointly for each column of sensordevices 801. The second source/drain terminal 804 b of the detectiontransistor 804 is coupled to a row line 809, which is provided jointlyfor each row of sensor devices 801. The changeover element 807 can beswitched in one of two switch positions “a” and “b”, depending onwhether a calibration mode or a measurement mode is to be set. Inaccordance with the scenario shown in FIG. 8, the switch element 807 isin the position “b”, so that the second source/drain terminal 806 b ofthe selection transistor 806 is coupled to a calibrationconstant-current source 810. By contrast, if the changeover element 807is in the switch position “a” (not shown in the figure), then the secondsource/drain terminal 806 b of the selection transistor 806 is coupledto a detection constant-voltage source 811. The value of the electriccurrent flowing on the row line 809 can be detected by means of acurrent detection unit 812, i.e. an ammeter for example.

Furthermore, a selection terminal 813 is coupled to the first columnline 808. If an electrical signal having a logic value “1” is applied tothe selection terminal 813, then all the selection transistors 806 ofthe associated column of sensor devices 801 are electrically conductive,so that the associated column of sensor devices 801 c is selected.Furthermore, a calibration terminal 814 is in each case coupled to arespective second column line 815, a common second column line 815 and acommon calibration terminal 814 being provided for each column of sensordevices 801. If a signal having a logic value “1” is present at thecalibration terminal 814, then all the calibration transistors 805 ofthe associated column of sensor devices 801 are electrically conductive,thereby enabling calibration.

The functionality of the sensor arrangement 800 is described in moredetail below.

Firstly, a description is given of how the sensor devices 801 arecalibrated column by column in order to compensate for fluctuatingproperties between different sensor devices (for example fluctuations inthe value of the threshold voltage of the detection transistors 804). Inorder to calibrate the n-th column, the latter is activated as solecolumn by a control signal having a logic value “1” being applied to theselection terminal 813 of the n-th column of sensor devices 801. Asignal having a logic value “0” is present at the selection terminals813 of all the other columns of sensor devices 801. As a result, thechannel regions of all the selection transistors 806 of the n-th columnof sensor devices 801 are electrically conductive, whereas the selectiontransistors 806 of all the other columns of sensor devices 801 arenonconductive. A read-out or calibration circuit with the changeoverelement 807 is in each case situated outside each row of sensor devices801. In order to carry out the calibration, the switch is brought to theposition “b”. This scenario is shown in FIG. 8. For calibration, asignal having a logic value “1” is momentarily applied to thecalibration terminal 814 of the n-th column of sensor devices 801, sothat the calibration constant-current source 810 is coupled to the gateterminal 804 c of the detection transistor 804 via the conductiveselection transistor 806 and the conductive calibration transistor 805.As a result, a constant current I_(kal) is impressed into the sensordevice 801. If the calibration transistor 805 is in the on state, thenexactly the electrical voltage required to derive the electric currentI_(kal) through the detection transistor 804 is established at the gateterminal 804 c of the detection transistor 804. This voltage isdifferent for each sensor device 801 of the n-th column of sensordevices 801 since the electrical parameters of the detection transistor804 may vary on account of statistical effects. If the coupling of thegate terminal 804 c and the first source/drain terminal 804 a of thedetection transistor 804 via the calibration transistor 805 isinterrupted again by the signal at the calibration terminal 814 beingbrought to a logic value “0”, clearly the electrical gate voltageassociated with the electric current value I_(kal) is stored at the gateterminal 804 c of the detection transistor 804. This calibration methodis gradually repeated for all the columns.

A measurement phase of the sensor arrangement 800 is described below.

For this purpose, the changeover element 807 is switched to the switchposition “a” (not shown in FIG. 8). As a result, the constant voltageV_(drain) is impressed into all the circuit devices 801 of an associatedrow of circuit devices 801 using the detection constant-voltage source811. The sensor arrangement 800 is sequentially read column by column. Acolumn to be read is selected by the associated selection terminal 813being brought to an electrical potential with a logic value “1”, so thatall the selection transistors 806 of the associated column of sensorarrangements 801 are brought to an electrically conductive state. If nosensor event takes place at a sensor element of a sensor device 801 of aselected column of sensor devices 801, then that DC component that wasstored on the gate terminal 804 c of the detection transistor 804 in thecalibration phase flows through the activated sensor device 801.Parameter fluctuations, in particular of the detection transistors 804,are therefore compensated for. In other words, the output signal for anidentical sensor event is identical and does not depend on thefluctuating parameters of the transistors. If a sensor event gives riseto a modulation of the electrical potential on a sensor element, thenthis results in a modulation of the electrical voltage at the gateterminal 804 c of the associated detection transistor 804 andconsequently of the electric current at the first source/drain terminal804 a of the detection transistor 804. This modulation is detected bymeans of the current detection unit 812 and can be amplified by externalamplifier elements. The electric sensor current signal may optionally beconverted into an electrical voltage (cf. evaluation unit 700 from FIG.7).

To summarize, it shall be emphasized that the sensor devices 801 can beactivated and deactivated column by column by means of the sensorarrangement 800. A sensor signal is amplified or converted into anelectric current. Statistical fluctuations of parameters of the sensordevices 801 are compensated for by means of calibrating the sensordevices 801.

A description is given below, referring to FIG. 9, of a sensorarrangement 900 in accordance with a second preferred exemplaryembodiment of the invention. The sensor arrangement 900 shown in FIG. 9is a modification of the sensor arrangement 800 shown in FIG. 8.Identical or similar components are therefore provided with the samereference numerals.

The essential difference between the sensor arrangement 900 and thesensor arrangement 800 is that the calibration transistor 805 isconnected up in a modified manner in the case of the sensor devices 901of the sensor arrangement 900.

In the case of the sensor arrangement 900, the second source/drainterminal 805 b of the calibration transistor 805 is coupled to thesecond source/drain terminal 806 b of the selection transistor 806 anddirectly to the row line 809. By virtue of the calibration transistor805 of the sensor arrangement 900 being directly coupled to the row line809 (clearly read-out line), it is possible to compensate for slightfluctuations which may arise via the selection transistor 806. However,the parasitic capacitance at the output node of a sensor arrangement 901in accordance with the sensor arrangement 900 may be somewhat highersince, in this case, two transistors are permanently connected by theirsource/drain terminals to the common output node of a sensor arrangement901 of a row.

A description is given below, referring to FIG. 10, of a sensorarrangement 1000 in accordance with a third preferred exemplaryembodiment of the invention. Those components of the sensor arrangement1000 which also occur in the sensor arrangement 800 or 900 are providedwith the same reference numerals.

The sensor arrangement 1000 is a matrix-type arrangement of amultiplicity of sensor devices 1005.

A sensor element of a sensor device 1005 is again symbolized as voltagesource 802, which, in accordance with FIG. 10, are connected between aterminal at electrical ground potential and the capacitor 803. Thecapacitor 803 represents a dielectric layer on the detection transistor804, by means of which the sensor element 802 is decoupled from thedetection transistor 804. The capacitor 803 is coupled to the firstsource/drain terminal 1001 a of a switching transistor 1001. The secondsource/drain terminal 1001 b of the switching transistor 1001 is coupledto the first source/drain terminal 805 a of the calibration transistor805 and to the gate terminal 804 c of the detection transistor 804.Furthermore, the gate terminal 1001 c of the switching transistor 1001is coupled to a switching terminal 1002 via a third column line 1003. Aseparate third column line 1003 is provided for each column of sensordevices 1005. The first source/drain terminal 804 a of the detectiontransistor 804 is coupled to the first source/drain terminal 806 a ofthe selection transistor 806, the gate terminal 806 b of which iscoupled to the selection terminal 813 via the first column line 808. Thegate terminal 805 c of the calibration transistor 805 is coupled to thecalibration terminal 814 via the second column line 815. The secondsource/drain terminal 805 b of the calibration transistor 805 is coupledto a first row line 1006, which is provided jointly for each row ofsensor devices 1005. The first row line 1006 is coupled to a calibrationconstant-voltage source 1004. The second source/drain terminal 806 b ofthe selection transistor 806 is coupled via a second row line 1007 tothe detection constant-voltage source 811, which is in turn coupled tothe current detection unit 812. The current detection unit 812 iscoupled to the calibration constant-voltage source 1004.

The functionality of the sensor arrangement 1000 is described in moredetail below.

In the case of the sensor arrangement 1000, the sensor devices 1005 canbe activated and deactivated column by column. The sensor signal or areference signal can be amplified or converted into an electric current.Furthermore, the sensor device 1000 is suitable for a correlated doublesampling for the purpose of eliminating parameter fluctuations.

It should be noted that the n-MOS transistors 1001, 805, 806 coupled toone of the column lines 1003, 815 and 808, respectively, can be broughtto an electrically conductive state by application of an electricalsignal having a logic value “1” to the associated terminal 1002, 814 and813, respectively, and thus represent a negligibly small nonreactiveresistance. By contrast, if the electrical signal at an associatedterminal 1002, 814 and 813, respectively, is at a logic value “0”, thenthe driven transistor is in the off state, in which case the leakagecurrents of the field-effect transistors can be disregarded.

The sensor arrangement 1000 is based on the correlated double samplingprinciple (CDS), whereby parameter fluctuations and low-frequency noiseare suppressed. In accordance with the CDS method, it is customaryfirstly to apply a reference signal to an input of an amplifier. Theamplified reference signal plus an offset signal of the amplifier isthen stored at the output of the amplifier. In a next phase, the sensorsignal is applied to the amplifier. The amplified measurement signalincluding the offset signal is then present at the output of theamplifier. Forming the difference between the two values makes itpossible to eliminate the offset signal of the amplifier.

The sensor devices 1005 of the sensor arrangement 1000 are sequentiallyread column by column. A column (e.g. the n-th column) of sensor devices1005 is in each case activated by the selection terminal 813, the firstcolumn line 808 and consequently the gate terminal 806 c of theselection transistors 806 of the associated column being brought to alogic value “1”. The pixels of a column are read in two phases inaccordance with the exemplary embodiment.

In a first phase, the reference voltage V_(kal) of the calibrationconstant-voltage source 1004, in an associated sensor arrangement 1005,is converted into an electric current and the value thereof is detected.For this purpose, an electrical signal having a logic value “1” isapplied to the calibration terminal 814, so that the calibrationtransistors 805 coupled thereto via the second column line 815 arebrought to an electrically conductive state. By contrast, the switchingterminal 1002 is at a logic value “0” in this phase, so that theswitching transistor 1001 is electrically nonconductive. The referencevoltage V_(kal) is present at the gate terminal 804 c of the detectiontransistor 804, which results in an associated electric current throughthe first source/drain terminal 804 a of the detection transistor 804.The value of said electric current may be different for different sensordevices 1005 of the sensor arrangement 1000 on account of statisticfluctuations of transistor parameters. The value of said electriccurrent is detected and stored in the read-out circuit of the respectiverow as electric reference current I_(meas)(m).

The actual sensor signal is detected in a second phase. In a shortestpossible time interval with respect to the first phase, the electricalsignal at the calibration terminal 814 is brought to a logic value “0”as a result of which the calibration transistors 805 turn off. Bycontrast, an electrical signal having a logic value “1” is applied tothe switching terminal 1002, so that the switching transistors 1001 arebrought to an electrically conductive state. As a result, a change inthe electrical potential at the sensor element 802 is mapped on the gateterminal 804 c of the detection transistor 804, which leads to amodulation of the electric current through the first source/drainterminal 804 a of the detection transistor 804. The value of saidelectric current is detected, and the difference between the detectedcurrent values from the first and the second phase is then formed. As aresult, parameter fluctuations between the different sensor devices aresuppressed and the output signal obtained depends exclusively on thesensor event.

The following publication is cited in this document:

[1] Stevanovic, N., Hillebrand, M., Hosticka, B. J., Teuner, A. (2000)“A CMOS Image Sensor for High-Speed Imaging”, IEEE InternationalSolid-State Circuits Conference 2000:104-105

LIST OF REFERENCE SYMBOLS

-   100 Sensor arrangement-   101 First sensor device-   102 Second sensor device-   103 Third sensor device-   104 Fourth sensor device-   105 Multiplexer-   106 Read-out circuit-   200 Sensor arrangement-   201 Nonreactive resistance-   202 Capacitance-   300 Diagram-   301 Abscissa-   302 Ordinate-   303 Signal profile curve-   400 Output driver circuit-   401 MOS transistor-   402 Voltage source-   403 Constant-current source-   404 Nonreactive resistance-   405 Capacitance-   406 Supply voltage-   407 Ground potential-   410 Equivalent circuit diagram-   411 Controlled current source-   412 Internal resistance-   500 Output driver circuit-   510 Equivalent circuit diagram-   511 Controlled current source-   512 Internal resistance-   600 Sensor device-   601 Field-effect transistor-   601 a First source/drain terminal-   601 b Second source/drain terminal-   601 c Gate terminal-   602 Voltage source-   603 Ammeter-   604 Supply voltage-   605 Ground potential-   606 Capacitor-   700 Evaluation unit-   701 Impedance-   702 Input-   703 Operational amplifier-   703 a Inverting input-   703 b Noninverting input-   703 c Output-   704 Constant-voltage source-   800 Sensor arrangement-   801 Sensor device-   802 Voltage source-   803 Capacitor-   804 Detection transistor-   804 a First source/drain terminal-   804 b Second source/drain terminal-   804 c Gate terminal-   805 Calibration transistor-   805 a First source/drain terminal-   805 b Second source/drain terminal-   805 c Gate terminal-   806 Selection transistor-   806 a First source/drain terminal-   806 b Second source/drain terminal-   806 c Gate terminal-   807 Changeover element-   808 First column line-   809 Row line-   810 Calibration constant-current source-   811 Detection constant-voltage source-   812 Current detection unit-   813 Selection terminal-   814 Calibration terminal-   815 Second column line-   900 Sensor arrangement-   901 Sensor device-   1000 Sensor arrangement-   1001 Switching transistor-   1001 a first source/drain terminal-   1001 b Second source/drain terminal-   1001 c Gate terminal-   1002 Switching terminal-   1003 Third column line-   1004 Calibration constant-voltage source-   1005 Sensor device-   1006 First row line-   1007 second row line

1. A sensor arrangement having a plurality of sensor devices formed atleast one of on or in a substrate, each of the sensor devices having: anelectrical signal converter; a sensor element coupled to the signalconverter, in which the sensor element can be used to characteristicallyinfluence the electrical conductivity of the signal converter on accountof a sensor event on the sensor element; a device for keeping constantan electrical voltage present at the signal converter; and a device fordetecting the value of the electric current flowing through the signalconverter as a sensor signal, wherein the electrical signal converter isa field-effect transistor having a gate terminal coupled to the sensorelement, the device for keeping constant an electrical voltage keepsconstant the electrical voltage between source and drain terminals ofthe field-effect transistor; and the sensor arrangement furthercomprises a calibration device for calibrating a respective sensordevice, the calibration device usable to bring a gate region of thefield-effect transistor to an electrical calibration potential such thatthe electric current is independent of parameter fluctuations of thefield-effect transistor. 2-3. (canceled)
 4. The sensor arrangement asclaimed in claim 1, further comprising an evaluation unit, which isprovided with the value of the electric current as sensor signal.
 5. Thesensor arrangement as claimed in claim 4, in which the evaluation unitforms, from the value of the electric current, an electrical voltagecharacteristic of the value or maps the value of the electric currentonto a digitally coded value that characterizes the value of theelectric current.
 6. The sensor arrangement as claimed in claim 5, inwhich the evaluation unit has an operational amplifier comprising: afirst input, to which the sensor signal can be applied; a second input,to which an electrical reference potential can be applied; and anoutput, at which the characteristic electrical voltage is provided; thefirst input and the output being coupled to one another by means of anonreactive resistor.
 7. The sensor arrangement as claimed in claim 1,configured as a biosensor arrangement.
 8. (canceled)
 9. The sensorarrangement as claimed in claim 1, in which the calibration device isset up such that an electric calibration current can be applied to thegate terminal and to one of the source and drain terminal of thefield-effect transistor for calibration purposes.
 10. The sensorarrangement as claimed in claim 4, in which the evaluation unit has acorrelated double sampling device, which forms, in the case of a sensorevent, a value of the electric current that is independent of parameterfluctuations of the field-effect transistor.
 11. The sensor arrangementas claimed in claim 10, in which the correlated double sampling deviceis set up such that, by means of the correlated double sampling device:in a calibration phase, a gate region of the field-effect transistor isbrought to an electrical calibration potential and the associated valueof the electric current is detected as a calibration signal and stored;in a detection phase, the value of the electric current on account of asensor event is detected as a sensor signal; in an evaluation phase, thesensor signal and the calibration signal are evaluated jointly.
 12. Thesensor arrangement as claimed in claim 1, in which the sensor devicesare arranged essentially in matrix form at least one of on or in thesubstrate and are connected up by means of row and column lines suchthat the sensor devices can be driven individually, row by row or columnby column.
 13. The sensor arrangement as claimed in claim 12, in whichat least one evaluation unit, at least one of: at least one calibrationdevice or at least one correlated double sampling device is providedjointly for at least a portion of the sensor devices of a row line or acolumn line.
 14. A method for operating a sensor arrangement: with asensor arrangement having a plurality of sensor devices formed at leastone of on or in a substrate, each of the sensor devices having: anelectrical signal converter; a sensor element coupled to the signalconverter, in which the sensor element can be used to characteristicallyinfluence the electrical conductivity of the signal converter on accountof a sensor event on the sensor element; a device for keeping constantan electrical voltage present at the signal converter; a device fordetecting the value of the electric current flowing through the signalconverter as a sensor signal, wherein the electrical signal converter isa field effect transistor having a gate terminal coupled to the sensorelement, the device for keeping constant an electrical voltage keepsconstant the electrical voltage between source and drain terminals ofthe field-effect transistor; in which case, in accordance with themethod, the electrical conductivity of the signal converter ischaracteristically influenced on account of a sensor event on the sensorelement; the electrical voltage at the signal converter is keptconstant; the electric current flowing through the signal converter isdetected as sensor signal; and in which at least a portion of the sensordevices is calibrated by a gate region of the respective field-effecttransistor being brought to an electrical calibration potential suchthat the value of the electric current in the case of a sensor event isindependent of parameter fluctuations of the field-effect transistor.15-16. (canceled)
 17. The method as claimed in claim 14, in which avalue of the electric current that is independent of parameterfluctuations of the field-effect transistor is formed using a correlateddouble sampling method in the case of a sensor event.